Low-noise high efficiency bias generation circuits and method

ABSTRACT

A bias generation method or apparatus defined by any one or any practical combination of numerous features that contribute to low noise and/or high efficiency biasing, including: having a charge pump control clock output with a waveform having limited harmonic content or distortion compared to a sine wave; having a ring oscillator to generating a charge pump clock that includes inverters current limited by cascode devices and achieves substantially rail-to-rail output amplitude; having a differential ring oscillator with optional startup and/or phase locking features to produce two phase outputs suitably matched and in adequate phase opposition; having a ring oscillator of less than five stages generating a charge pump clock; capacitively coupling the clock output(s) to some or all of the charge transfer capacitor switches; biasing an FET, which is capacitively coupled to a drive signal, to a bias voltage via an “active bias resistor” circuit that conducts between output terminals only during portions of a waveform appearing between the terminals, and/or wherein the bias voltage is generated by switching a small capacitance at cycles of said waveform. A charge pump for the bias generation may include a regulating feedback loop including an OTA that is also suitable for other uses, the OTA having a ratio-control input that controls a current mirror ratio in a differential amplifier over a continuous range, and optionally has differential outputs including an inverting output produced by a second differential amplifier that optionally includes a variable ratio current mirror controlled by the same ratio-control input. The ratio-control input may therefore control a common mode voltage of the differential outputs of the OTA. A control loop around the OTA may be configured to control the ratio of one or more variable ratio current mirrors, which may particularly control the output common mode voltage, and may control it such that the inverting output level tracks the non-inverting output level to cause the amplifier to function as a high-gain integrator.

This patent application is a national stage application filed pursuantto 35 U.S.C. §371 of international application number PCT/US2009/004149filed Jul. 17, 2009 (published Jan. 21, 2010 as publication numberWO/2010/008586 A2), which application claims priority to U.S.application No. 61/135,279 filed Jul. 18, 2008 and entitled “Low NoiseCharge Pump with Common-Mode Tuning Op Amp”, the entire contents ofwhich are hereby incorporated by reference herein, and herebyincorporates by reference the entire contents of U.S. patent applicationSer. No. 10/658,154 filed Sep. 8, 2003 and entitled “Low Noise ChargePump Method and Apparatus”, issued May 18, 2010 as U.S. Pat. No.7,719,343. The above-identified U.S. provisional patent application No.61/135,279 filed Jul. 18, 2008 and international application numberPCT/US2009/004149 filed Jul. 17, 2009 are hereby incorporated byreference herein in their entirety.

BACKGROUND

1. Field

The present disclosure is widely applicable to electronic integratedcircuits (“ICs”).

2. Related Art

It is usually desirable for an IC to operate from a single voltagesupply. However, many ICs require two or more different voltagesupplies, for example to provide internal bias supplies, for idealoperation. Such different supplies can be provided externally to theintegrated circuit, but this is undesirable from a user standpoint.Providing additional supplies is not only inconvenient for the user, butmay also cause the conductors coupling such external supplies to the ICto be unduly long, which among other difficulties may cause undesiredemissions if noise is present on the supply. As such, it is a commonpractice to provide auxiliary circuitry on ICs to generate suchadditional bias generation voltages, or other voltage supplies, as maybe required for circuit operation needed. Charge pumps are one of themost common of such auxiliary voltage-generating circuits used in ICs.

However, charge pumps have characteristics that have rendered themdifficult to use in certain applications. In particular, charge pumpshave invariably created a substantial amount of electrical noise.Regulations have been promulgated to prevent electronic devices frominterfering with each other, and such regulations establish maximums forallowable emissions. In some applications the noise generated by acharge pump may cause the IC or system in which such IC is disposed toexceed such maximum permitted noise emissions.

For example, most radios, cell phones, TVs, and related equipment todayrequire an “RF switch” to control connections between varioustransmitter and receiver circuits (“RF” is used generically herein tomean any reasonably high frequency ac signal). At least one auxiliaryvoltage generator is often needed to satisfactorily bias the FETs thatcomprise a semiconductor RF switch. Many of the products that employ RFswitches are transceivers, such as cell phones, that are subject tostringent regulatory limitations on the electrical signals that it ispermitted to emit. Because such RF switches are directly connected tothe transceiver antenna, even very small amplitude noise signalsgenerated by the bias generator of the RF switch will be all tooefficiently radiated. It has been found that the noise generated by aconventional charge pump may be sufficient to cause a cell phoneemploying an RF switch using such charge pump to exceed the maximumnoise emissions permitted by applicable regulations. As such, a noisycharge pump can render such a cell phone unsuitable for its commercialpurpose.

Consequently, bias generation circuits, such as charge pumps thatgenerate far less noise than conventional charge pumps are crucial forcertain applications. Low noise bias generation circuits will findadvantageous employment in a wide range of integrated circuits, whetherto satisfy regulatory spurious emission limits, or to avoid interferencewith other local circuitry. Such circuits must also be efficient interms of integrated circuit area consumed, and, especially for batteryoperated devices such as cellular telephones, must be efficient in termsof power consumption.

Additionally, it is often useful to control an output voltage of acharge pump by means of a feedback control loop that includes adifferential-input operational transconductance amplifier (“OTA”). OTAoutput common mode voltages include the effects of various offsets,including input signal misalignment, differential input offset voltages,finite gain of input common mode signals, and other mismatches that mayoccur throughout the OTA. Nulling the effect of such offsets isparticularly useful for amplifying small signals. Adjusting outputvoltage levels is also useful for permitting maximum gain before thesignal is clipped.

The method and apparatus presented herein address the need forlow-noise, high efficiency bias generation circuits, including chargepumps, regulation control and amplification circuits, bias level settingcircuits and, particularly for the capacitive coupling of low-noiseclocking waveforms, efficient active bias circuits. Various aspects ofthe bias generation method and apparatus described herein will be seento provide further advantages, as well.

SUMMARY

A bias generation method and apparatus is set forth that may generatebias voltage supplies quietly and efficiently by means of a charge pumpthat alternately couples charge from an input supply to a transfercapacitor and then couples the charge to an output, and may couple biasvoltages to nodes requiring biasing by means of “active bias resistor”circuits. A variety of novel features are described and employed toachieve such bias generation. Many charge pump topologies are possible,some of which are set forth in U.S. patent application Ser. No.10/658,154, which is incorporated by reference; many charge pump clockoscillators are suitable, especially that produce waveforms havinglimited harmonic content above a fundamental frequency, which may besubstantially sine-like, and which moreover may include two waveformssubstantially symmetric and in phase opposition. Such charge pump clocksmay be coupled to transfer coupling switch control nodes viacapacitance, and the nodes may be biased to selected levels by means ofcharge conducted by active bias resistors that may not have anysubstantial resistance at all. Moreover, the bias voltage generation maybe controlled by an amplifier loop that includes an operationalamplifier circuit having a controllable current mirror ratio, which maypermit common mode control of differential outputs from the amplifier.

One important aspect of the bias generation circuits and method is afocus on minimizing the extent to which a charge pump creating biasvoltages generates and transfers electrical noise to nearby circuits anddevices with which the charge pump is associated. Some features of thebias generation circuits and method aid in reducing such noisegeneration and conduction, while others serve to permit bias generationto be efficient in terms of integrated circuit area and powerconsumption while employing such noise reduction features. Any one ormore of these various features may be combined in bias generationcircuits and methods that generate reduced interference.

Because the clock that controls a charge pump or other clocked biasgeneration circuits is both a direct and an indirect source ofundesirable electrical noise currents, characteristics of the clockdefine some embodiments of the bias generation circuits and method.Embodiments may be defined by the clock they employ to control switchingdevices in a charge pump, according to any combination of one or more ofthe following features, each of which contribute to low noise generationin a charge pump. Because it is desirable for the output to have lowharmonic content, one distinguishing feature of an embodiment of acharge pump is a clock with an output having low harmonic content asdefined by any of the specific harmonic content limits set forth herein.As harmonic content of a clock output is reduced it generally becomesmore sinusoidal, so such a clock output may be defined as substantiallysine-like. Alternatively, the harmonic power divided by the power at thefundamental frequency fo (i.e., total harmonic distortion “THD”), may belimited to not more than −5 dB, or −10 dB, or −20 dB, or even −30 dB. Asa further alternative, such a clock output may be defined as restrictedto having third harmonic power that is less than −20 dB, −30 dB, or −40dB compared to the power at fo. The clock waveform may also be describedas containing amplitudes of each harmonic of the fundamental frequencythat decrease by at least 20 dB per decade, or by at least 30 dB perdecade, or at least 40 dB per decade. Thus, for a waveform having afundamental operating frequency fo of 8 MHz and an amplitude A₁ for its8 MHz sinusoidal wave component, the amplitude A_(N) of every harmonicsinusoidal component at frequency N*fo, N an integer, may be required tobe no greater than A₁ reduced by 20, 30 or 40 dB/decade, i.e., A_(N)(dBA)≦A₁ (dBA)−2*N, or A_(N) (dBA)≦A₁ (dBA)−3*N, or A_(N) (dBA)≦A₁(dBA)−4*N. Which of these varying quality levels is required for theclock waveform will typically depend upon the problem, based on aparticular hardware implementation in combination with desired emissionlimits, which the embodiments described herein are employed to solve.

Low harmonic content signals are not readily produced, or reproduced, bydigital circuits, which leads to several features that distinguish anddefine embodiments of a bias generation method or apparatus withreference to their controlling clocks. Embodiments of certain chargepumps may be defined by having their clock output(s) capacitivelycoupled to most or all transfer capacitor switches they control, whichis rendered advantageous for suitable low-harmonic clock signals due tothe limitations of digital circuits. Also, particularly because suitableclock waveforms typically drive a switch into conduction with only halfof the peak-to-peak amplitude, it is important that the waveform belarge compared to the supply available to generate it. Suitable clockwaveforms may be required to have a peak-to-peak amplitude that is atleast 95%, 98% or 99% of the amplitude of a supply from which such clockis generated.

As an aid to biasing capacitively coupled control signals to thetransfer capacitor switching devices, it may be helpful to employ activebias “resistors,” active circuits that couple a bias voltage on a firstnode to a second node coupled to a transistor control node. A goal is tocouple the first node bias voltage to the second node without undulyreducing the amplitude of an alternating drive signal also applied tothe second node, which drive signal may be oscillating and may moreoverbe substantially sine-like. Embodiments of such active bias-couplingcircuits may be configured to substantially reduce, compared to thevoltage between the two nodes, a voltage appearing across an impedancelimiting current between the two nodes, or alternatively may entirelyavoid the presence of significant resistors and limit current conductionby capacitive charging, with further current through active devices asmay be suitable. They may also substantially preclude current fromflowing between the first and second nodes when a voltage therebetweenis small compared to peak voltages between the nodes, small beingdefined as less than about 0.4V, or about 0.8V, or about 1.2V, or beingalternatively defined as less than about 25%, or about 50%, or about 70%of the peak voltages. Embodiments may further comprise a capacitiveelement to charge to a portion of the peak voltage between the first andsecond nodes, and may be a bridge circuit whereby alternating polarityvoltage between the two nodes causes a varying but unipolar voltageacross a current limiting circuit. The current limiting circuitcomprises a series circuit of a resistor of less than about 10 MΩ, acapacitance that may be shorted, and an active current limiting circuitthat may be shorted. The current limiting series circuit may be disposedin parallel with a bypass circuit to conduct non-linearly greatercurrents for node voltages that exceed a selected voltage.

The transfer capacitor switching devices each have a correspondingthreshold voltage Vth at which they begin conducting, and in generalshould be biased to a voltage related to such Vth. To provide areference for such bias voltage without unduly absorbing supply current,a switched-capacitor bias supply circuit is described. Embodimentsdischarge (or charge) a capacitive element during first periodicportions of a clock signal, and charge (or discharge) it through adiode-connected device during second periodic portions of the clocksignal while the device is coupled to an output storage capacitor. Suchbias supplies may use a single clock signal, or may use a plurality ofclock signals, which may differ from each other in phase relationshipand/or average voltage level. Such bias supplies may be particularlyconfigured to function with sine-like clock signals.

Because producing two clock phases that are matched and have appropriatecharacteristics is difficult, a further separate definition of a chargepump clock is having two phases generated by a ring oscillator that mayhave any odd or even number of inverter stages coupled in a ring,including at least one differential inverter stage. Differentialinverter stages may each have “first” and “second” inverter sections,and all “first” inverter sections may be sequentially coupled in thering and all “second” inverter sections also sequentially coupled in thering, except that ring oscillators employing an even numbers of invertersections will cross-couple the outputs of one inverter section bycoupling its “first” output to a sequentially next inverter section“second” input, and its “second” output to the sequentially nextinverter section “first” input. Ring oscillators having an even numberof inverter stages may also be required to include a startup circuit.Such a startup circuit may sense a non-oscillation condition, or moreparticularly a common mode stage output condition, and thereafter mayprovide an oscillation drive signal, which may more particularly be adrive that separates output voltages of one of the differential inverterstage outputs. Differential ring oscillators having an odd number ofinverter stages may be further required to include a phase lockingcircuit to ensure differential phasing; an appropriate phase lockingcircuit may include two additional inverter stages coupled inanti-parallel between the inputs and outputs of an otherwise ordinarydifferential inverter stage, or else a pair of capacitors similarlycross-coupled. Whether a ring oscillator is differential or not, it maybe advantageous for producing low harmonic content output(s) to limitthe number of inverter stages, because fewer stages tend to produce adesirably less square output waveform; consequently, a charge pump clockmay be defined as limited to 2, 3 or 4 inverter stages.

A control circuit may be employed to regulate a charge pump outputvoltage to a desired value. An amplifier circuit is needed for suchcontrol, and accordingly embodiments of an operational transconductanceamplifier (OTA) are described, but the OTAs are suitable for generalapplication. A differential amplifier circuit (OTA-diff amp) of the OTAhas differential inputs coupled to the transistor control nodes in adifferential pair of transistors having common drains coupled to ashared current source circuit. The OTA-diff amp includes an additionalvariable ratio current mirror input node, a signal applied to whichsubstantially controls a ratio between sensed current in the drainbranch of one of the differential input pair transistors, and currentmirroring such sensed current in the drain branch of the othertransistor of the differential input pair. The variable ratio currentmirror input may be used, for example, to affect a gain of the OTA-diffamp, or for controlling an output voltage level. A differential outputOTA may have two variable ratio current mirror OTA-diff amps sharingopposite input nodes, and further may control the variable currentmirror ratio for each of the OTA-diff amps from a single common modecontrol input. An independent loop driving the common mode control inputmay be configured to control common mode voltage levels of the twooutputs of the differential output OTA to a selectable level, or maycause voltage levels of one of the two outputs to track voltage levelsof the other output, which may nullify effects of input misalignmentand/or may enhance gain of one of the OTA-diff amps.

Embodiments of the bias generation method or apparatus may employ anycombination of individual aspects of the method or apparatus, and may beemployed in a wide range of bias generation architectures andconfigurations.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments of the present invention will be more readily understood byreference to the following figures, in which like reference numbers anddesignations indicate like elements.

FIG. 1 is a simplified block diagram of a charge pump circuit configuredto produce a regulated output voltage that is higher than or opposite inpolarity to a source voltage.

FIG. 2A is a highly simplified representation of a charge pump in whicha charge pump clock conducts substantial current through a transfercapacitor.

FIG. 2B illustrates that a charge pump clock as shown in FIG. 2A may beseen as comprising a separate pre-clock that does not conductsignificant current through a transfer capacitor in combination withseparate transfer capacitor switching circuits.

FIG. 3 represents an architecture by which an exemplary embodimentproduces an output voltage of approximately negative two times a supplyvoltage.

FIG. 4 is a more detailed block diagram showing circuit elements of anexemplary charge pump.

FIGS. 5A, 5B and 5C schematically illustrate three active bias couplingcircuits.

FIGS. 6A, 6B, 6C and 6D schematically represent circuits for producing abias voltage while consuming reduced current from the supply, includingrepresentations for producing two different bias voltages using a singleclock phase, and representations for producing the two different biasvoltages using two related clock signals.

FIG. 7 is a block diagram of an exemplary high gain amplifier circuitfor use in controlling an output voltage of a charge pump.

FIG. 8 schematically represents a circuit for selectably limiting arange of an amplifier output signal.

FIG. 9 schematically illustrates schematically represents a differentialoutput operational transconductance amplifier having two differentialpair amplifier circuits that each include variable ratio current mirrorcircuit controlled by a same ratio control input voltage.

FIG. 10 is a simplified diagram demonstrating an alternative use for acommon mode tunable operational transconductance amplifier.

FIG. 11 is a simplified diagram demonstrating an alternativeimplementation of a variable ratio current mirror circuit.

FIGS. 12A and 12B schematically illustrate two exemplary current limiteddifferential inverter stages for a ring oscillator, the stage having ananti-parallel locking circuit to ensure phase opposition thedifferential outputs.

FIG. 13 schematically illustrates a current limited differentialinverter stage of a ring oscillator and optional cross-coupledcapacitors to ensure output phase opposition of the stage.

FIG. 14 is a simplified diagram representing features of a four-stagering oscillator coupled to a startup circuit.

DETAILED DESCRIPTION

The bias generation circuits described herein are fabricated onintegrated circuits, providing bias and other supply voltages. A biasgeneration method or apparatus may employ one or more charge pumps todevelop bias voltages. Charge pumps, as that term is used herein, aredefined by a process of storing charge from an input supply on atransfer capacitor, then switching the nodes to which the transfercapacitor is coupled so as to transfer some portion of that storedcharge to an output supply. The charge pumps described are expected tobe entirely within a single monolithic integrated circuit, exceptperhaps for filtering components such as external capacitors.

As was developed in the related U.S. patent application Ser. No.10/658,154 filed Sep. 8, 2003, entitled “Low Noise Charge Pump Methodand Apparatus” and incorporated herein by reference, a sinusoidal orsine-like charge pump clock output can reduce harmonic noise generationby the charge pump, particularly if the sine-like clock output is itselfgenerated in a manner that generates little harmonic noise. Capacitivecoupling of the clock signal may be useful to ensure that such an analogwaveform can control the variety of switching devices necessary to pumpcharge from a source supply to a different output supply, while ensuringthat simultaneous conduction is avoided for all switching devices thatare disposed in series across supply rails (or other low impedancenodes). In order for such coupling to work as intended, it is helpful tobias the switching devices very precisely, so that conduction to andfrom the transfer capacitors (sometimes called “fly” capacitors) canoccur for as much of the available time as possible, and with switchimpedance as low as possible.

These desirable conditions are best met by generating a charge pumpoutput that is substantially sine-like, or at least has limited harmoniccontent compared to a sine wave, and which has an amplitude as large aspossible in view of the voltage capabilities of the semiconductorprocess by which the integrated circuit containing such a charge pump isfabricated. Bias generation, more generally, may additionally requireprecise bias levels to be generated and conveyed to the switches withoutconsuming excessive power or integrated circuit area.

Bias supplies often need to be regulated, for which purpose case lowcurrent, high gain amplifiers are useful. The amplifiers used in anexemplary quiet, regulated charge pump have unusual features that are ofwide applicability. Examples are described herein of various aspects ofquiet, efficient bias generation.

Note that “output supplies” or “additional supplies”, or even “voltagesupplies” all refer to pairs of nodes within a circuit, a supply nodeand a reference node. The circuit creating such supplies are generallydesigned to maintain the difference between the node pair at a constant,DC (direct current) or zero frequency voltage. Other circuits willtypically interact with such supplies, causing them to have variations.However, except in the case of a variable-output supply, such variationsin voltage are incidental, and will be attenuated with respect to thesource of such changing signals. Supply voltages are designed to remainsubstantially constant under conditions of varying loads placed upon thevoltage, and their success at this function is often a primary measureof their quality. Variable-output supply voltages may be varied in valuefrom time to time under control of a control level, but even then willbe expected to remain substantially constant for times in excess ofseconds, and to be changed only according to operating conditioncircumstances. Voltages that are periodically changed to substantiallydifferent levels to turn on and off one or more circuit elements are notsupplies, or supply voltages, but are control signals or controlvoltages. This distinction, well understood though it is by most personsof skill in the art, is explained here because it is occasionallymisapprehended.

Overview

FIG. 1 illustrates the basic topology of an exemplary embodiment of thesubject bias generation circuits and method with a block diagram thatidentifies functional blocks of a pre-regulated charge pump. Apreregulation block 1 is connected to a reference node 2 and a supplynode 3 of a source supply 4. Under control of a feedback signal 6, apower control circuit 5 provides a controlled supply 8 that becomes theinput to a switched transfer capacitor circuit 10. The switched transfercapacitor circuit 10 includes one or more transfer capacitors 11 thatare coupled via switches, as represented by switch blocks 12 and 13, tocouple during some time periods to source connections 14 and 15, and tocouple during other time periods to output connections 16 and 17. Theremay be a plurality of transfer capacitors 11, and the source (14, 15)and output (16, 17) to which the transfer capacitors are coupled may beintermediate sources or outputs that are only indirectly related to theregulated source connections 8, 2 or the regulated output connections 18and 19. The switch blocks 12 and 13 represent any number of electronicswitches, such as FETs, that serve to connect the terminals of thetransfer capacitor(s) as necessary. These switches are controlled byswitch control circuits 20 via connections 22, under control of anoutput signal 31 of a clock generator 30.

Feedback circuitry 50 compares the output supply 18 (with respect to theoutput reference 19) to a reference voltage provided by a voltagereference 40. The feedback circuitry 50 produces a control signal thatcontrols the power control circuit 5. In the exemplary embodiment, apremium is placed on keeping voltages within the switched transfercapacitor circuits to a minimum. Pre-regulation ensures that the chargepump source Vcp 8 coupled to the switched transfer capacitor circuit 10will be no greater than necessary to provide the desired output voltage.As an alternative to such a pre-regulation topology, a regulationelement similar to pre-regulator 1 may be disposed after the switchedtransfer capacitor circuit 10 but before the output supply connections18 and 19, and may be similarly controlled by control signal 6.

Transfer Capacitor Switching Topology

Before turning to an exemplary topology, a distinction between twoclasses of such topologies is noted. A first class of charge pumptopologies are designated “control only clock” topologies, and aredistinguished by the fact that they do not transfer significant currentbetween the charge pump clock(s) and any transfer capacitors controlledby such clocks. A second class of charge pump topologies are designated“current transfer clock” topologies, and are distinguished by the factthat they include a charge pump clock output that is a primary source ofcurrent ultimately conveyed to the output via the transfer capacitor(s).

FIG. 1 is a block diagram of an example of a “control only clock”topology charge pump. Switches are arrayed around the transfercapacitor(s) 11 as needed (e.g., as represented by switch blocks 12 and13). Such “transfer capacitor switches” couple charge into and out ofthe transfer capacitors, from a source or to an output. The switchcontrol circuits 20, and in particular the output 31 of the charge pumpclock 30 that drives such control circuits, provide only control signalsto the transfer capacitor switches (12, 13). While some finite currentwill likely be conducted from the clock output 31 and control circuits20 into the transfer capacitor(s) 11, any such current is notsubstantial, but is merely incidental to providing control. For example,currents due to parasitic gate capacitances if the switches arecomprised of FETs, or base currents if the switches include bipolartransistors, may enter the transfer capacitors incidentally, but are notsignificant compared to the currents that the transfer capacitorswitches intentionally conduct into and out of the transfer capacitor(s)11.

FIG. 2A is a block diagram illustrating a simple example of “currenttransfer clock” topology charge pump, in which an output 32 of thecharge pump clock 3000 is coupled to a terminal 34 of a transfercapacitor 11 so as to supply substantial current into the transfercapacitor 11, and ultimately to the output supply 26 for storage on asmoothing capacitor 28. Transfer capacitor coupling switches, such asrepresented in a switch block 24, may be controlled by the charge pumpclock by means of switch control circuitry 20 via connections 22.However, the transfer capacitor coupling switches may also be devices,such as diode-connected FETs, that can be controlled by the charge pumpclock via the transfer capacitor 11 without a need for direct controllines 22. In any event, the distinguishing characteristic of this classof charge pump topologies is that the output from a charge pump clock3000 directly provides substantial current to a transfer capacitor 11.

The distinction between these two classes of charge pumps must beunderstood to avoid confusion when comparing different charge pumps, andparticularly different charge pump clocks. However, in some sense thedistinction is largely a matter of drawing convention. FIG. 2Billustrates blocks internal to the charge pump clock 3000 of FIG. 2A,including switches 3012 that couple the nominal clock output 32 toeither a source connection Vs1 3014 or a source connection Vs2 3015,under control of an output 3031 of a “pre-clock” block 3030. Thus, atleast in such instances, observation of clock design details permits thecharge pump of FIG. 2A to be seen as a “control only clock” topologythat has simply been drawn in a manner that lumps important switchingfeatures into a block labeled “clock.” In particular, the pre-clock 3030may be seen to be a “control only” charge pump clock having an output3031 that controls switches 3012 but does not conduct current directlyto a transfer capacitor (connected to 32, but not shown). Thus, currentto the transfer capacitor, via 32, comes not from the preclock 3030, butfrom sources Vs1 3014 and Vs2 3015 under control only of the pre-clock3030. Nonetheless, many charge pump references omit the details of thecharge pump clock output drive circuitry that would permit suchrecharacterization. Consequently, when comparing charge pumps asdescribed in different references, it is important to bear in mind thedistinctions between “control only clock” topologies and “currenttransfer clock” topologies.

An exemplary transfer capacitor switching topology for providing adoubled and inverted output is illustrated in the block diagram in FIG.3. The transfer capacitor coupling switches are represented by switchblocks 302, 304, 306 and 308, all of which may be considered toalternate between position A (during even time slots) and position B(during odd time slots). Thus, during an even time slot switch blocks302 and 304 are in position A, so a first transfer capacitor TC1 310 ischarged to a voltage Vcp by being coupled between a source connection312 (Vcp) and source connection 314 (0V). During the next odd time slotall four switch blocks go to position B. The positive terminal of TC1310 is coupled to 0V and the negative terminal to an intermediate pointVint 316, which after sufficient cycles will therefore be driven closeto −Vcp. During the same odd time slot, a second transfer capacitor TC2318 will be coupled by switch blocks 306 and 308 between sourceconnection 312 (Vcp) and the intermediate voltage Vint 316. Aftersufficient cycles, presuming the load is not excessive, TC2 318 willtherefore be charged to nearly 2*Vcp. During even time slots TC2 318 iscoupled between output 314 (0V, the same as the source connection) andoutput 320 (Vout). If the current drawn from output 320 (Vout) is notexcessive, after sufficient cycles Vout will approach −2*Vcp. Thetopology of the charge pump block diagramed in FIG. 3 is of the first“control only clock” type: the charge pump clock does not providesignificant current to a transfer capacitor, but instead provides onlycontrol signals to transfer capacitor switches.

FIG. 4 is a schematic block diagram illustrating some details of theexemplary transfer capacitor switching circuit. In general, the transfercapacitors TC1 310 and TC2 318 are switched in the manner shown in FIG.3. However, a number of the detailed circuit features of FIG. 4 areunusual. The clock output is provided in the form of two oppositephases, ø1 and ø2, at about 8 MHz. To reduce noise generation andtransmission, these clock signals should have limited harmonic contentas described elsewhere herein. To achieve limited harmonic content, thewaveform(s) should at least have well-rounded edges. More ideally, thelimited harmonic content will cause the waveform to be substantiallysine-like. It is also desirable, for the purpose of driving FETs as hardas possible for high efficiency, for the waveforms to have peak-to-peakamplitudes as large as practical in view of available voltages and thevoltage withstand capacity between terminals of the FETs. In anexemplary embodiment, the clock output amplitude is about 2.4V (peak topeak).

The details of this circuit are specific to the semiconductor processmost often used by the applicant, but the skilled person will have notrouble adjusting details to fit different semiconductor processingparameters. The process includes the following FET types, from which amajority of circuit components are fabricated. N-channel FETs include:Regular N (“RN”) FETs that have a nominal threshold voltage of 450 mV,High doping N (“HN”) FETs that have a nominal threshold voltage of 700mV, and Thick oxide High doping N (THN) FETs that have a nominalthreshold voltage of 900 mV. THN FETs have gate withstand voltages thatare about 3.6V, compared to the 2.7V withstand of RN and HN FETs. ADepletion-mode N (“DN”) is similar to HN and RN except for having athreshold voltage of about −1V, so that it is fully conducting underordinary circumstances. It has a standard gate voltage withstandcapacity. Corresponding P-channel FETs include RP (Regular), HP (Highdoping) and THP (Thick oxide High doping) P FETs having −400, −600 and−800 mV nominal threshold voltages, respectively. IN or intrinsic FETsmay have a threshold voltage of approximately 0 V.

Most capacitors are fabricated by connecting the drain and source of aDN FET as one terminal, and using the gate as the other terminal. Suchcapacitors have a working voltage only equal to the standard gatevoltage withstand capacity. Capacitance is reduced when an FET capacitoris biased off, which occurs when a DN FET with source and drain tiedtogether (i.e., capacitor configured) is charged such that the gate isabout 1V more negative than the channel. Therefore, adjustments must bemade for large signal bipolar operation. For example,metal-insulator-metal (“MIM”) capacitors may be used if linearity iscrucial, or two DN FETs may be disposed in anti-parallel if linearity isnot a concern. Capacitors formed of IN FETs, on the other hand, haveextremely non-linear characteristics: when the voltage goes to zero,i.e. to the threshold voltage, the channel becomes substantiallyineffective at creating a plate of the capacitor, and thus thecapacitance goes to a very low value of perhaps 20% of the capacitanceat higher voltages.

The transfer capacitor switching circuit of FIG. 4 has four voltagesupply rails: a voltage Vcp 312 that is controllable from 1.7V to 2.4V;0V 314; an intermediate voltage Vint 316 that is approximately −Vcpdepending upon load; and Vout 320, which is approximately −2*Vcpdepending upon load. As is described elsewhere, Vout will be controlledto have a magnitude of at least 3.4V (negative) by means of a feedbackloop that controls Vcp to be greater than about 1.7V, as needed, basedon a source of about 2.4V.

One terminal of TC1 310 is alternately coupled to either Vcp 312 via HP402, a P-channel FET, or to 0V 314 via HN 404, an N-channel FET. Bothdevices are driven from the same clock signal ø1, capacitively coupledto the gates of the devices via coupling capacitors C 406 and C 408,respectively. Cs 406 and 408 may be fabricated as DN FETs having a gatearea about 23 times larger than the gate area of their correspondingFETs HP 402 and HN 404, respectively. Because the devices are conductingonly half of the time, the effective capacitance of C 406 and C 408 isroughly 46 times larger than the effective capacitance between the gateand source of the corresponding FETs they are driving. The capacitanceof Cs 406 and 408 may be roughly 0.75 pF.

In the exemplary embodiment, the effective gate parasitic capacitance ofHP 402 is about 1/46 that of coupling capacitor C 406. Capacitivevoltage division would therefore attenuate the signal by about 2%, andthe signal on the gate of HP 402 would be about 98% of the clockvoltage. However, a gate bias voltage must be coupled to the gate, suchas via the bias impedance Z 412. If Z 412 is a resistance; it shoulddesirably be about 4 MΩ so as not to significantly further attenuate thegate drive signal. Depending upon the gain of the FET switches and theavailable magnitude of the clock signals ø1 and ø2, this may not be anissue. Depending upon process factors, it will be satisfactory in someembodiments to employ linear impedances to limit attenuation of theclock signal at the FET gates, at the clock operating frequency fo, thatis less than 20%, 10%, 5%, or 3%. The bias impedance Z 412 may be aresistor, or may have inductive characteristics to achieve sufficientlylow attenuation at fo.

Each of the other FET switches is also driven by either ø1 or ø2 in asimilar manner, so the gain values and the values of their correspondingcoupling capacitances and bias impedances will be selected according tothe same considerations described with respect to HP 402. Typically, ø1and ø2 will have the same amplitude, each HP 402, 414, 416 and 418 willhave substantially similar characteristics, so each correspondingcoupling capacitor C 406, C 420, C 422 and C 424 will have the samevalue, as will each corresponding bias impedance Z 412, Z 426, Z 428 andZ 430 and each bias voltage RP_Vt 410, 432 and 434.

Similarly as the P-channel FETs, the N-channel FET switches HN 404, 436,438 and 440 will generally also have substantially identicalcharacteristics as each other. Consequently, the correspondingcomponents, including bias impedances Z 442, Z 444, Z 446 and Z 448,coupling capacitors C 408, C 450, C 452 and C 454, and bias voltagesHN_Vt 456, HN_Vt 458 and HN_Vt 460, may also be identical to each other.

However, coupling ø2 to HP 416 and HN 438 requires higher voltagecapacitors than the usual 3V DN FET capacitors of the exemplarysemiconductor fabrication process. Accordingly, these low-voltage DN FETcapacitors are disposed in series to increase the effective voltagewithstand capability. Due to the circuit configuration, C 422 and C452are made 2 times as large as other coupling capacitors, such as C 406,and C 462 is made 4 times as large, so that the effective amplitude onthe gates of HP 416 and HN 438 is approximately the same as on each ofthe other transfer capacitor switch FET gates. The junction between C462 and Cs 422, 452 is biased to a midpoint voltage by coupling it toRP_Vt 432 via bias impedance Z 464.

Transfer capacitor TC1 may be about 15 to 30 pF, while TC2 may also be15 to 30 pF. Larger transfer capacitors increase efficiency, but mayrequire a large semiconductor area. In the exemplary embodiment, TC1 andTC2 are fabricated as capacitor-connected DN FETs having a workingvoltage of only about 2.7V. The voltage stresses on TC2 exceed thewithstand capacity of single devices, so TC2 is actually fabricatedusing two capacitances in series. To obtain a given capacitance, the ˜6VTC2 therefore requires 4 times greater area than the ˜3V TC1. In view ofthis area penalty, TC1 may be made relatively large (e.g., about 30 pF)while TC2 is left at 15 pF and thus is only twice the size. If aparticular fabrication process has no area penalty, it would be moreeffective to increase both devices by about the same factor.

It is important to avoid simultaneous conduction for transfer capacitorswitch pairs disposed across a low impedance source, such as HP 402 andHN 404 disposed across Vcp 312 and 0V 314. To this end, both devices areturned off when the clock drive signal is between its average value and200 mV below that value. The average or bias voltage on the gate of HP402 is controlled by an RP_Vt tracking source 410 as coupled to the gatevia a large bias impedance Z 412. However, the threshold voltagemagnitude of HP devices is about 200 mV larger (˜−600 mV) than for RPdevices, so the RP_Vt tracking source 410 sets the bias voltage about200 mV smaller (˜−400 mV) than the threshold voltage of the HP FETs suchas HP 402 (˜−600 mV). The N FETs, however, such as HN 404, are biased totheir threshold voltage of about 700 mV by an HN_Vt bias supply (HN_Vt456, for HN 404). Thus, both devices in each FET pair are biased “off”for about 200 mV out of the range of the clock signal, which in the 1.2V peak waveform of the exemplary clock signals is equivalent to an offtime of slightly over 5% of the clock half-period, or about 3.3 ns whenfo=8 MHz. This small nominal off time is adequate because variations ofparameters between physically close devices within these integratedcircuits will be very small, and will tend to track each other acrossoperating conditions.

Active Bias “Resistor”

In the exemplary charge pump circuit it is desired to maintain maximumclock signal amplitude on the gates of the transfer capacitor switchingFETs. To avoid attenuating the gate signal amplitude, the magnitude ofthe bias impedances will need to be quite large, ideally about 4 MΩ inthe exemplary embodiment. In some semiconductor processes a simpleresistor of such magnitude may take more device area, and/or theresulting impedance may be more difficult to control, than the impedanceof an extensive active circuit. An active bias impedance circuit mayhave a complex impedance that includes a significant inductivecomponent. However, an active circuit that ensures the correct biasvoltage on a capacitively-coupled FET gate presented with a uniformoscillating signal need not present a linear impedance at all. Instead,a completely nonlinear active circuit may be employed as an “active biasresistor” circuit.

FIG. 5A schematically illustrates one example of such an active biasresistor circuit disposed between terminals A 502 and B 504. It is ahighly nonlinear bridge circuit that conducts very little current insteady-state operation. Presuming that an oscillating voltage ofsufficient magnitude appears between terminals A 502 and B 504, thecircuit conducts current as necessary to equalize the magnitudes of thealternating peak voltages between the A and B terminals. When the peakvoltages are equal, the midpoint voltages will also be equal. In theexemplary embodiment one of the terminal voltages is a DC value (Vt), sothe circuit serves to ensure that the positive and negative peaks of theother terminal are precisely balanced about the DC value.

Let Vabp be the magnitude of the peak voltage during “positive” halfcycles when the voltage on terminal A 502 is greater than the voltage onterminal B 504, and let Vbap be the magnitude of the peak voltagesduring the other “negative” half cycles when the voltage of terminal B504 is greater than the voltage of terminal A. When the voltage ofpositive half cycles exceeds the threshold of THN 508 (about 900 mV),FETs THP 506 and THN 508 are turned on, coupling the series connected C510 and R 512 across A and B (R 512 coupled to terminal A 502). C 510may be a capacitor-connected DN FET of about 0.5 pF, and R 510 is about93.5 kΩ, so they form a series RC circuit establishing a pole at about3.4 MHz. During each positive half cycle C 510 will conduct current thatreflects an average value of Vab while Vab>900 mV. The same happensduring the negative half cycle when the voltage of terminal B exceedsthat of terminal A by 900 mV; except THP 514 and THN 516 conduct whenVba exceeds 0.9V (THP 506 and THN 508 are off). Therefore, terminal B isconnected to R 512, so while Vba>900 mV, C 510 conducts current thatreflects the average value of Vba during this period.

If the average value of Vab (while Vab>0.9) is greater than the averagevalue of Vba (while Vba>0.9), more current flows from terminal A toterminal B during positive half cycles to raise the voltage on C 510.Presuming that Vba is still less than Vab, current flows out of C 510during negative half cycles, causing a net negative current fromterminal B to terminal A, equivalent to a net positive current from A toB. Thus, during each half cycle a net current moves from thehigher-voltage terminal (the terminal having the higher average voltageduring the period it is more than 0.9V greater than the lower terminal)and into the lower-voltage terminal (the terminal having the loweraverage voltage during the period that it is more than 0.9V greater thanthe higher-voltage terminal). Given the lack of DC current through thegate-connected terminal, this will force the two terminals to experienceidentical average voltages during that portion of their respective halfcycles when the bridge is conducting (i.e., V>0.9V). For any waveformsbetween terminals A and B having positive and negative half-cycles thatare symmetrical, forcing the average voltages during the conductingperiods to be equal is equivalent to forcing the overall average valueof each terminal voltage to be equal, thus equalizing the DC or biasvoltage on the two terminals.

Current does not flow at all when the voltage across terminals A and Bis less than about 900 mV. During the time that threshold is overcome, C510 tries to charge to the average voltage during both half cycles.Presuming that the peak voltages between terminals A and B are 1.2 V andthat the threshold of THNs 508 and 516 is 0.9V, current will flow forabout the middle 80 degrees of each half wave, and C 510 will charge toabout 1.1 V. Average currents over each half cycle will of course bezero at steady state, but the current that does flow will be nonsinusoidal. FETs THN 518 and THN 520, which have a nominal combinedforward voltage of 1.8 V, permit rapid bias adjustments by enablinglarge currents to flow when one of the peak voltages, nominally 1.2 V,reaches 1.8 V.

FIG. 5B illustrates a simple alternative circuit for setting the bias ona gate in the presence of an oscillating drive signal. When the voltageon terminal A 502 exceeds the voltage on terminal B 504 by the combinedthreshold voltage magnitudes for diode-connected RN 522 (˜450 MV) and RP524 (˜400 mV), current flows from A to B as limited by R 526 (e.g., 200kΩ). When the voltage on terminal B 504 exceeds that on terminal A 502by the combined threshold voltage magnitudes for diode-connected RN 528and RP 530, current flows from B to A as limited by R 526. Thus, netcurrent flows from the terminal experiencing the greater averagepositive voltage excursions with respect to the other terminal. Thethreshold combinations of the two anti-parallel diode-connected FETpairs will be well matched, so the average current flow during the twohalf cycles will balance when the average voltages are balanced.

FIG. 5C illustrates a further alternative circuit for setting the biason a gate in the presence of an oscillating drive signal. Unlike FIGS.5A and 5B, FIG. 5C is completely symmetric with respect to terminals A502 and B 504, two identical circuits being disposed between theseterminals in anti-parallel. Also, the circuit of FIG. 5C does notrequire any large resistor at all. Instead, current limiting is achievedby a “switched capacitor” effect: the current depends on charging anddischarging a small capacitor on each cycle of the input signal.

For each half cycle that V_(A) (voltage of A 502) exceeds V_(B) (voltageof B 504), capacitor C1 532 of only about 12.5° F. limits the chargecoupled from terminal A to B. The charge would be entirely capacitivedisplacement current through either FET 536 (during the positive halfcycle) or FET 534 (during the negative half cycle), and thus of zeroaverage value, were it not for the current through FETs 538 and 540,which mirror only the positive half-cycle current throughdiode-connected FET 536. FET 540 is not essential, but mitigates Vdschannel modulation of mirror FET 538. For the opposite half cycle whenV_(B) exceeds V_(A), the anti-parallel circuit consisting of C2 542 andFETs 544, 546, 548 and 550 perform inverse-identically as C1 532 andFETs 534, 536, 538 and 540, respectively.

There are particular features of the circuit that are useful, but notessential. As one example, C1 532 is an FET configured as a capacitor,with source and drain coupled together. In an exemplary embodiment, theFET is an “INA” type, which indicates that it is “intrinsic”, N-channel,and of size A (channel width 1.4 microns and length 2 microns, generallyindicated as W/L=1.4/2 microns). In the particular process, the IN-typeFET has a threshold voltage Vth of approximately zero volts. When Vgs(=Vgd) is less than Vth (zero), the channel of the FET practicallydisappears, so that the capacitance of C1 532 is only about 20% as largeas when Vgs is greater than zero. In the exemplary embodiment described,C1 532 has a capacitance of approximately 12.5 fF when Vgs>0V, but onlyabout 2.5 fF when Vgs<0V. One typical AC voltage across the terminals Aand B 502 and 504 is 1.2 V peak, and the RN-type FETs have Vth of about0.7 V. During recovery, when V_(B) exceeds V_(A), C1 532 eventuallysupports a negative voltage across its terminals of about −0.5 V.However, the amount of charge required to establish −0.5 V in thiscondition is only about 1/5 as much as would be required to establish+0.5V. As a result, C1 exits the “reset” half cycle with very littlecharge that would otherwise require displacement current when V_(A)exceeded about 0.2V; the current is negligible until V_(A) rises toabout 0.6V. This helps keep the overall charge transfer per cycle small,so that the active bias circuit has the low current consumption of avery large value resistor. However, it requires far less integratedcircuit area than such a large value resistor.

The circuit of FIG. 5C functions best when the amplitude of theoscillating waveform imposed across terminals A and B 502 and 504 has apeak value greater than Vth of FETs 536 and 538 (and the Vth of FETs 546and 548 for the opposite half cycle). However, the only upper limit onthe amplitude of the oscillating waveform across A and B is thebreakdown voltage of C1 532 and FET 540 (C2 542 and FET 550 for theopposite half cycle), increased by the smallest threshold voltage Vth ofFETs 534 and 536 (the smallest Vth of FETs 544 and 546 for the oppositehalf cycle).

The charge that is conducted by the active bias resistor illustrated inFIG. 5C on the positive half cycle is the displacement current of C1532, plus the mirrored current in FET 538. The displacement current willflow back during the negative half cycle to reset C1 532, leaving themirrored current in FET 538 as the net charge coupled from A 502 to B504. On the opposite half cycle, the net current is the mirrored currentthrough FET 548. To the extent that the threshold voltages are matched,and that the capacitance of C1 532 is equal to the capacitance of C2542, the charge coupled from A to B will only become zero when thehalf-cycle amplitudes are identical. Otherwise, net current will flow,which will move the average voltage of the biased node, e.g., V_(A),toward the bias source voltage, e.g., V_(B). The biased node istypically the gate of a relatively large FET.

Active bias resistors, like ordinary bias resistors, cause the voltageof a selected node to go to the same average value as that of a biasvoltage applied to one side of the circuit. However, active biasresistors may cause the node voltage to reach the bias levelsignificantly more quickly than would an ordinary resistor thatperformed the same function and conducted the same average current. Anembodiment such as illustrated in FIG. 5C may be particularly suited forsuch faster tracking. However, although resistors are not required forthe embodiment illustrated in FIG. 5, resistors may be employed withsome embodiments of this circuit. Any such resistors may, however, belimited to maximum values that do not exceed 100 k ohms, or 50 k ohms,or 20 k ohms, 10 k ohms or even 1 k ohms. Very small resistors may beused without penalty because they require very little integrated circuitreal estate, but larger resistors occupy substantial space.

Vt Tracker

A threshold-setting circuit may simply be a diode-connected FET coupledto a source voltage via a limiting resistance. However, to minimize thesource load for battery-powered devices, the limiting resistance mayneed to be very large, thereby operating the diode-connected FET atextremely low current, and also imposing an area penalty for the largeresistance in many semiconductor processes. Accordingly, FIGS. 6A-Dschematically illustrate switched-capacitor circuits for providing biasvoltages for FETs while drawing very little charge from the source (lowaverage current), where the clock(s) may be substantially sine-like.FIGS. 6A and 6B show Vt trackers that use a single clock phase toprovide Vt for HN and RP FETs respectively. FIGS. 6C and 6D show Vttrackers that use two clock phases ø1 and ø2 to provide more robust Vttrackers for HN and RP FETs, respectively.

In FIG. 6A a source VDD 602 (less than 2.5V) with respect to VSS 604begins to charge C 606 (4 fF) via THN 608 and HN 610 when the clockvoltage rises to about 1.6 V (Vt THN ˜900 mV, Vt HN ˜700 mV). At aboutthe same time, THN 612 turns on, coupling the output HN_Vt 614 andsmoothing capacitor C 616 (200 fF) to C 606 and diode-connected HN 610,which sets the output level. C 606 charges to a voltage of (VDD−HN_Vt),providing current if needed to C 616. As the clock (2.4 V p-p) passesthe 2.4 V peak value and returns to about 1.6 V, THNs 612 and 608 and HN610 turn off, and THP 618 turns on, discharging C 606 and turning offTHNs 608 and 612 more forcefully. This condition prevails until theclock passes the negative peak value of about 0 V and increases to about1.6 V, at which point another cycle begins. C 606 and C 616 may becapacitor-coupled DN FETs of appropriate area.

FIG. 6B is the RP-FET analog of FIG. 6A, but Vdd 601 should be less than2.1 V. When the clock signal declines from its peak (about 2.4V) toabout 1.2 V below Vdd, which may be substantially sine-like, (i.e., lessthan 0.9V), C 620 (4 ff) begins to charge via THP 622 (Vt ˜800 mV) andRP 624 (Vt ˜400 mV). THP 626 turns on thereafter, coupling the drain ofdiode-connected RP 624, which sets the output voltage level, to theoutput RP_Vt 628, and the smoothing capacitor C 630, via THP 622. Whenthe clock signal returns to 1.2 V below Vdd 601, RP 624, THP 622 and THP626 turn off. THN 632 turns on when the clock signal reaches ˜0.9V,discharging C 620. THN 632 should not turn on significantly before RP624, THP 622 and THP 626 turn off, for which reason Vdd 601 should notexceed 2.1 V.

FIG. 6C schematically illustrates a switched capacitor HN_Vt trackerthat employs a clock phase at two different bias points to render thecircuit more tolerant of parameter variations. The supply VDD 602/VSS604 charges C 606 (4 fF) via RN 634 and HN 610, and the Vt-setting drainvoltage of diode-connected HN 610 is coupled to the output HN_Vt 614 andthe smoothing capacitor C 616 (˜200 fF) via RN 634 and RN 636, when theclock signal clk_n 638 exceeds about 1.15 V. Clk_n 638 may be anapproximate sinusoid of about 2.4 V p-p, biased to have an averagevoltage of HN_Vt (about 700 mV) above VSS 604. Accordingly, clk_n 638exceeds 1.15 V for only about the middle 136 degrees of its 180 degreepositive half cycle, leaving about 22 degrees of non-conduction at eachend. HN 610 sets the output level for HN_Vt at about 700 mV, and thethreshold of RNs 634 and 636 are about 450 mV. Clk_p 640 issubstantially identical to clk_n 638 except that it is biased to anaverage voltage of RP_Vt, about 400 mV below VDD 602. When clock signalclk_p 640 is more than 0.4 V below VDD 602, RNs 634 and 636, and HN 610,must be off, as RP 642 is on to discharge C 606. This condition willexist for almost exactly the full negative half cycle of the clockwaveform. Note that clk_n 638 may be capacitively coupled to a clockoutput, and may be biased by disposing an active bias “resistor,” alsodescribed elsewhere herein, between clk_n 638 and HN_Vt 614. Similarly,clk_p 640 may be capacitively coupled to the same clock output byanother capacitor, and biased by disposing an active bias “resistor”between clk_p 640 and the output RP_Vt 648 of FIG. 6D. Clk_p 640 andclk_n 638 are not heavily loaded, so the same signals may be sharedbetween the circuits of FIGS. 6C and 6D.

FIG. 6D schematically illustrates an RP Vt tracking circuit generallyconverse to that of FIG. 6C, and may use the same two clock signalsclk_n 638 (biased to HN_Vt) and clk_p 640 (biased to RP_Vt with respectto VDD 602). During that portion of the clock negative half cycle whenclk_p 640 is more than about 0.8 V below VDD 602 (0.4 V below the biaslevel), C 620 (4 fF) charges via threshold-setting diode-connected RP624 and RP 644, and RP 646 couples the drain voltage of RP 624 to theoutput RP_Vt 648 and smoothing capacitor 630. During the positive clockhalf cycle, when clk_n 638 is greater than its bias level HN_Vt (about0.7V), HN 650 is turned on to discharge C 620. HN 650 is on forapproximately the entire clock positive half cycle, but does not conductconcurrently with RPs 624 and 644, and RP 646 which are on only when theclock is about 400 mV or more below its bias point, leaving about 20degrees of nonconduction at each end of the clock signal negative halfcycle.

Charge Pump Output Control Feedback Circuit Details

Block 50 of FIG. 1 is an integrating amplifier that compares the outputs18 and 19 from the charge pump to a reference voltage provided by block40, generating from any error a voltage 6 to control a preregulationcircuit 5. Any good differential input operational amplifier may be usedfor block 50, but the exemplary embodiment employs some unique circuitryfor this function, as outlined in FIG. 7, particularly in a differentialCommon Mode controlling Operational Transconductance Amplifier(“CM_OTA”).

The overall integrating amplifier 50 of FIG. 7 includes a differentialCM_OTA 710 having differential inputs 712 and 714, normal and invertingdifferential outputs 716 and 718 respectively, and a CM_tune input 720that reduces the output common mode voltage between outputs 716 and 718when the CM_tune input voltage is increased. The positive output 716 ofthe differential CM_OTA 710 provides the output drive signal to controlthe pre-regulator (not shown here). The gain of the differential CM_OTA710 is controlled by an internal common-mode feedback loop within theintegrating amplifier 50.

The common-mode feedback loop drives the CM_tune input 720 as necessaryto adjust the common-mode voltage of the differential output of theCM_OTA 710, such that negative differential output 718 has the sameaverage value as the positive differential output 716. A unity gainbuffer 730 provides a current-limited replica of the positive output 716of the CM_OTA 710 to a range limiter 740. A simple single-ended OTA 750is configured as an amplifier 760 that integrates differences betweenthe inverting output 718 of the CM_OTA and the range-limited version 742of normal output 716. The gain magnitude of amplifier 760 is limited to0.5 by R 762 (200 kΩ) and 764 (100 kΩ) above a frequency, set by C 766(300 fF), of around 5 MHz, which is somewhat lower than the operatingfrequency fo (around 8 MHz) of the charge pump this circuit serves. Inthe exemplary embodiment, each of the outputs of amplifiers 710, 730 and750 has a current drive capacity of less than 2 μA. Note that a smallcapacitor (˜100 fF, not shown) between the inverting output 718 of theCM_OTA 710 and ground may be useful, in view of the limited currentcapacity, to reduce high frequency loop gain for added stability.

The output range limiter 740 is often needed because the integrator 50is designed to operate with high gain, such that an input differentialof more than a few 10 s of mV can saturate and lock up the feedback.FIG. 8 schematically illustrates a suitable range limiting circuit. Therange-limited output signal 742 is prevented from going more positivethan the voltage at upper limit 802 plus the forward voltage of D 804,and is prevented from going more negative than the voltage at lowerlimit 806 minus the forward voltage of D 808. Ds 804 and 808 may bediode-connected FETs, for example RP FETs having a forward voltage ofabout 400 mV. RN 810 (Vth about 450 mV) sets a current of about 1.5 μAbased on a current-setting voltage biasn1, with the drain voltage of RN810 controlled to a level set by biasn2 816 by cascode configuration ofRN 814. RP 818 and RP 820 are diode connected in the exemplaryembodiment, conducting all of the current set by RN 810 if the voltageof upper limit 802 is less than about 800 mV below VDD. The signal at742 cannot exceed the forward voltage of D 804 above that voltage, or inother words the positive excursion of signal 742 is limited to aboutVDD-400 mV.

The signal 742 is similarly limited to be not lower than the forwardvoltage of D 808 (about 400 mV) below the lower limit voltage 806without sinking all the current provided by RP 822 based on biasp1 824with the drain voltage provided by cascode RP 826 as controlled bybiasp2 828. The higher of Vlow1 830, which is applied to the gate of RP832, and Vlow2 834, which is applied to the gate of RP 836, sets lowerlimit voltage 806. Signal 742 will be clipped if it drops low enough toforward bias D 808.

Common Mode Voltage Controllable Differential OTA

FIG. 9 schematically illustrates exemplary details of the differentialCM_OTA 710 of FIG. 7. The current for the positive differential inputpair FETs 902 and 904 is set by RN 906 in conjunction with cascode RN908 to less than 2 μA. FET 902 establishes current for current mirrorsensing device RP 910. The current mirror causes the current provided tothe drain of FET 904 to substantially reflect the current conducted byRP 910. However, a ratio between the current conducted through RP 910(the sensed current) and the current delivered to the output at thedrain of FET 904 (the mirrored current) may be continuously controlledby a voltage provided to a common mode control input CM_tune 912.Presuming that normal common mode feedback is enabled by CMF_on 914being held low, the fixed reflection ratio of about 1/2 provided by RP916 may be augmented by additional reflective conduction in RPs 918 and920. If CM_tune 912 is high enough to turn RP 922 completely off, RP 916is half the size and thus reflects about half the current sensed in(conducted by) RP 910, for a current mirror ratio of 2:1. However, ifCM_tune 912 is quite low, then RPs 916, 918 and 920 reflect the currentof RP 910 increased by a current mirror ratio of 1:2, because the totalarea of RPs 916, 918 and 920 is twice the area of RP 910. As CM_tune 912is decreased, RPs 918 and 920 reflect a progressively larger multiple ofthe current of sensing device RP 910. Thus, CM_tune can control theeffective current mirror ratio for differential input pair 902 and 904over a range from about 1:2 to about 2:1.

A differential amplifier circuit within an OTA (OTA-diff amp) is acircuit having inputs to each of an input differential pair oftransistors (such as 902 and 904) that are connected in common-emitteror common-source configuration. The common source is coupled to acircuit behaving roughly like a current source (e.g., properly biasedRNs 906 and 908). Such an OTA-diff amp has two branches, one coupled tothe drain or collector of each of the input differential pair devices.It is typical for one of the branches to conduct current through acurrent sensing element for a current mirror, as for example RP 910, andfor the other branch to receive “mirrored” current from a “mirroring”circuit that reflects the current conducted by the sensing element, suchas by having a comparable device biased to a gate voltage developed bythe sensing device. Typically the mirroring circuit is a single devicethat closely matches the sensing device and thus sets a mirror ratio ofabout 1:1. In circuits that are designated “variable ratio currentmirror OTA-diff amps,” however, the effective ratio between the sensedand mirrored current is not only not necessarily 1:1, it is madecontinuously variable on the basis of a control input. One way toachieve that is to control an effective size of the mirroring circuit ascompared to the sensing circuit. In FIG. 9, for example, devices 916,918 and 920 may all be part of the mirroring circuit (if RP 932 isbiased on). However, the drain voltage of RP 922 will affect aneffective contribution to such mirroring circuit by RPs 918 and 920;thus, controlling the drain voltage of RP 922 is capable of continuouslycontrolling the effective current mirror ratio. An alternative methodfor varying the current mirror ratio is to use a simple mirroringcircuit, such as a single FET, in one branch, but to controllablyparallel or shunt the current sensing circuit in the other branch,either changing the effective size of the sensing circuit, or elsechanging the proportion of branch current that is conducted, and thussensed, by a sensing device. This alternative is illustrated in FIG. 11.

Thus, a variable ratio current mirror OTA-diff amp includes anadditional variable ratio current mirror input node, a signal applied towhich substantially controls a ratio between current sensed in onebranch of the differential input pair transistors, and mirrored currentin the other branch that reflects the sensed current. The variable ratiocurrent mirror input in such an OTA-diff amp may be used, for example,to affect a gain of the OTA-diff amp, or for controlling an outputvoltage level taken from one of the branches. The circuit of FIG. 9, forexample, employs two different variable ratio current mirror OTA-diffamps to control a common-mode voltage of differential outputs.

In FIG. 9, the gates of FET 902 and FET 904 are the plus (in P) andinverting (MN) inputs, respectively, for the first differential inputpair. The gates of FETs 925 and 924 are the plus and inverting inputs,respectively, for a second differential input pair. A noninvertingoutput (outP) 926 of the CM_OTA is at the drain of FET 904 of the firstinput differential pair, while an inverting output (outN) 928 is at thedrain of FET 925 of the second input differential pair. RP 930 sensescurrent for the current mirror of the second differential amplifiercircuit.

RP 932 and RP 934, when turned off by a high voltage on CMF_on input914, serve in both differential pair circuits to prevent current throughthe largest of the mirroring FETs in both differential circuits (RPs 920and 940). In the second differential pair circuit, RPs 936, 938 and 940serve the same purpose as is served by RPs 916, 918 and 920 in the firstdifferential pair circuit. RNs 942 and 944 in the second differentialpair circuit also function the same as RNs 906 and 908 of the firstdifferential pair circuit. In the exemplary embodiment, RPs 916 and 918,as well as 936 and 938, are each half the effective size of thecorresponding current setting FETs, RPs 910 and 930, respectively. RPs920 and 940 are equal in size to RPs 910 and 930. Accordingly, if RPs932 and 934 are turned off but RP 922 is fully turned on, then eachcurrent mirror is fixed at a ratio of approximately 1:1. In theexemplary circuit, if CMF_on 914 is disabled (high), CM_tune can stillhave some effect on the current mirror ratio, and should be pulled fullylow to fix the current mirror ratios to about 1:1.

The variable current mirror ratio is controlled for both differentialinput pairs by RP 922. The current in the non-output branch of eachdifferential circuit (RPs 910 and 930) are fixed current sensors for therespective current mirrors, while the outputs 926 and 928 are connectedto the selectable FETs 918 and 920, and the selectable FETs 938 and 940,respectively. Accordingly, increasing the conduction of RP 922 raises avoltage level of both outputs: outP 926 and outN 928, raising the commonmode output voltage. The converse occurs when the conduction of RP 922is decreased; thus, RP 922 controls the common mode output voltage ofthe differential output CM_OTA of FIG. 9.

Referring again to the common mode control loop 50 of FIG. 7, the commonmode voltage of CM_OTA 710 is the midpoint between the average voltagesof the differential outputs 716 and 718. Because the positive output 716is coupled to the inverting connection of the amplifier 760, CM_tune 720will change inversely to positive output 716, so the output common modevoltage will follow the positive output 716. The common mode voltage isdriven to equal the dc level of positive output 716, which occurs whenthe dc level of the inverting output 718 is equal to the dc level of thepositive output 716. Because these two conditions are equivalent, it canbe accomplished by merely driving the common mode voltage untilinverting output 718 has the same average voltage as positive output716. Because the common mode control loop 50 causes the positive output716 to rise further, for a signal that causes it to rise initially, theloop increases the gain of the CM_OTA 710, particularly at lowerfrequencies. As a result, the CM_OTA 710 is able to function like anintegrator, having extremely large gain for low frequency input offsets.

Variable ratio current mirror OTA-diff amps are employed in the CM_OTA710 to increase gain in the amplifier, particularly at lowerfrequencies. However, a CM_OTA can be employed to set the differentialoutput common mode level to any desired level, as illustrated in FIG.10. A differential CM_OTA 710, as in FIG. 7, has positive and invertingoutputs 716 and 718. The common mode output voltage, established by Rs101 and 102, may be compared to an arbitrarily selected voltage 103(typically an output range midpoint) by a single ended OTA 750,configured as an integrator by C 104 with optional high frequency gainsetting resistor R 105. Many other configurations are possible.

FIG. 11 schematically illustrates an alternative configuration of avariable current mirror in a differential amplifier circuit. In FIG. 11a common mode control voltage CMCV 111 has an opposite polarity as doesCM_tune of FIG. 9, because the output common mode voltage will tend tofollow CMCV 111, whereas it tends in the opposite direction of CM_tune.HP 112 controls an effective “size” of the combination of RPs 113 and114, which is one way to vary the mirror ratio. It is also possible forRP 113 to simply siphon current around the bias setting FET RP 114, suchthat the mirrored current from RP 115 will reflect only a portion ofcurrent in the “+” branch of the diff amp. Depending upon theapplication, the RP 113 may, for example, be three times the size of RP114, while mirroring FET RP 115 may be twice the size of RP 114. An“enable” input may be added, and the ratio made fixable at 1:1, in themanner analogous to circuitry performing those functions in FIG. 9.

The variable ratio current mirror circuit of FIG. 11, as describedabove, may be used to replace the corresponding mirror components in adifferential CM_OTA such as illustrated in FIG. 9. However, because ofthe sense inversion of the CMCV input, if such a CM_OTA is employed asillustrated in FIG. 7, the polarity of the amplifier 760 will also beinverted.

FIG. 11 illustrates a simplified circuit, suitable for processes havingmodestly sized high value resistances, by means of which gain can beboosted for a single differential amplifier having a variable ratiocurrent mirror. Resistors 116 and 117 set a range for CMCV 111 as afunction of Vo, while they operate with R 118 and C 119 to roll off gainat high frequency as necessary for stability. Replacing HP 112 by ahigher threshold, lower gain THP FET may permit shorting R 116 andopening R 117 to reduce size requirements at the expense of gain. Inmany semiconductor processes it may be more practical to replace any orall of the components C 119 and Rs 116, 117 and 118 with activecomponents, and thereby to produce similar or better results.

Single ended differential amplifiers having a variable ratio currentmirror controlled by an input voltage may be suitable for many otherpurposes. For example, they may be used to nullify the effects of inputmisalignment or voltage offsets. They may also be employed as a thirdinput to independently modulate a signal amplified by the differentialamplifier circuit. The polarity of such an input may be selected byusing a variable ratio current mirror as in FIG. 9 or as in FIG. 11.

Low Noise Differential Charge Pump Clock

A sinusoidal (or sine-like) charge pump clock signal is very useful forcontrolling a charge pump without generating spurs and undesirableharmonic noise. However, there are some drawbacks to employingsinusoidal clock signals to drive switching devices: available clockoutput amplitude is difficult to employ to achieve ample drive levels,because if switching occurs near the waveform midpoint, then onlyapproximately half of the peak-to-peak waveform amplitude is availablefor driving a control node into its conduction voltage range. Employinga plurality of clock phases can simplify some charge pump designconsiderations, but will typically entail a need to accurately controlthe timing and/or amplitude relationships between the different clockoutput phases.

In general, making the clock outputs more sinusoidal reduces the amountof undesirable electrical noise that is generated. While perfect sinewave outputs are not possible, a waveform quality should be selectedthat provides adequate performance for the intended use of the circuit.The clock outputs may be required substantially sine-like, but adesigner can almost arbitrarily select how sine-like to make theoutputs; each improvement will result in a reduction of electrical noiseat some nodes or locations, but each improvement may incur an addedcost, such as in design effort and integrated circuit area usage.

Various parameters may be employed to define a clock waveform that issuitable to solve a particular noise problem created by bias generation,or other supply voltage generation, within an integrated circuit. Aparameter of total harmonic distortion of a clock output compared to aperfect sine wave at the operating frequency fo is defined as the sum ofthe power in all harmonics of fo contained in the waveform, divided bythe power in the fundamental frequency fo. Using that definition, indifferent embodiments the waveform may usefully be limited to having aTHD of no more than −5 dB, −10 dB, −20 dB, or −30 dB. In someapplications the third harmonic may be of particular interest, anddifferent embodiments may require the third harmonic power to be no morethan −20 dB, −30 dB, −40 dB or even −50 dB with respect to thefundamental power at fo. Also, it may be useful to control the clockwaveform such that the amplitude of each harmonic of the fundamentalfrequency is rolled off by at least 20, 30 or 40 dB/decade. Thus, for awaveform having a fundamental operating frequency fo of 8 MHz and anamplitude A₁ for its 8 MHz sinusoidal wave component, the amplitudeA_(N) of every harmonic sinusoidal component at frequency N*fo, N aninteger, may be required to be no greater than A₁ reduced by 20, 30 or40 dB/decade. That is, using dB or dBA units for each quantity, theharmonic amplitudes may need to be limited such that [A_(N)≦A₁−2*N], or[A_(N)≦A₁−3*N], or [A_(N)≦A₁−4*N]. Expanding the last for clarity:[A_(N) (dBA)≦A₁ (dBA)−4*N (dBA)]. Alternatively, the amplitudes of theharmonic components may be limited as follows (in volts):[A_(N)≦A₁/N/m], where depending upon circumstances m may need to beequal to 0.7, 1, 1.5, 2, 2.5, 3, 4 or 6. Which of these varying qualitylevels is required for the clock waveform will typically depend upon theproblem, based on a particular hardware implementation in combinationwith desired emission limits, which the embodiments described herein areemployed to solve.

Capacitive coupling of clock signals to control switching devices, whichare necessarily disposed at many different potentials within a chargepump circuit, is sufficiently convenient to justify the relatively largesemiconductor area required for adequate capacitors. However, capacitivecoupling of a sinusoidal signal generally entails driving a switchingdevice on with only half of the overall clock waveform (generally eitherthe positive or negative half cycle of a clock signal). Accordingly,when supply voltages are small, it is not easy to provide sufficientdrive voltage to the charge pump switching devices. Therefore, it willbe helpful to provide charge pump clock signals that not only have twoinverse sine-like phases, but that also have a peak-to-peak amplitudenearly equal to the available supply voltage.

Some exemplary embodiments of a low-noise charge pump clock employdifferential inverter stages. The differential stages may be designed toensure large amplitude signals that extend very nearly to the supplyrails, and of course provide complementary outputs. Low noise operationis facilitated by current limiting each inverter in each stage. Asubstantially sine-like output, or any output having very low harmoniccontent beyond the operating frequency fo, may more readily be generatedby employing less than five inverter stages in a ring oscillator. Adifferential ring oscillator has the advantage of permitting ringoscillators to have any number of stages, including both odd and evennumbers, contrary to ordinary ring oscillator teaching. Some embodimentsof the low noise charge pump clock may include a differential ringoscillator having two, three or four inverter stages.

A differential ring oscillator having an odd number of inverter stageswill oscillate unconditionally, but it is possible for the two invertersin an inverter stage to have a common-mode output: the same, rather thanan opposite, voltage at each moment on the two outputs. Differentialring oscillators having an odd number of stages, such as three, maytherefore benefit from a method of ensuring that the two inverters ofeach inverter stage are in opposite phase. Such phase-separatingcircuitry in a single inverter stage may be sufficient, but phasecontrol circuitry in other stages may also be helpful.

An exemplary design of a differential inverter stage that includes ananti-parallel inverter lock circuit is illustrated in FIG. 12A. THP 121and THN 122 form a basic inverter of the positive input in P 123 to thepositive inverted output outN 124, while THP 125 and THN 126 areconfigured as a complementary inverter from the inverted (or negative)input in N 127 to the inverted (and thus now positive) output outP 128.To limit the drive capacity of the inverters, and thus to slow andsmooth the output transitions, both inverters are coupled to VDD via acurrent limiting circuit comprised of RPs 129 and 130, and are coupledto GND via a current limiting circuit comprised of RNs 131 and 132. RP129 and RN 131 set the current based on bias voltages biasp1 and biasn1,respectively, while RP 130 and RN 132 are configured in cascodeconnection, biased by biasp2 and biasn2 respectively, to limit thecurrent source sensitivity to output voltage. Because current settingFET RP 129 operates at nearly zero drain voltage irrespective of thevoltage on the drain of RP 130, this cascode configured current sourceis capable of providing current even when the drain voltage of RP 130 isnearly VDD. Similarly, cascode configured RNs 131 and 132 are capable ofproviding correct current over all output voltages on the drain of RN132 to within a few mV of ground. Thus, the output waveform may be tunedto achieve a p-p voltage nearly equal to the supply voltage VDD withrespect to ground. These FETs 121-122, 125-126, and 129-132 constitute acomplete basic differential inverter stage having both positive andinverted inputs 123 and 127 and positive and inverted outputs 124 and128.

The remaining circuitry of FIG. 12A constitutes anti-parallel couplingthat may be incorporated in one or more inverter stages of adifferential oscillator having an odd number of stages. It is possiblefor the positive and inverted sections of a differential ring oscillatorwith an odd number of stages to operate at any phase relationship withrespect to each other, so some provision is needed to ensure theirphases are separated by 180 degrees. A first inverter comprised of THP133 and THN 134 is cross-coupled with a second inverter comprised of THP135 and THN 136. These FETs 133-136 may be made smaller, for example 70%as large, compared to the FETs 121-122 and 125-126 of the primaryinverters of the stage. More importantly, the current sources thatcouple these anti-parallel inverters to VDD and GND may be designed forfar less current than is provided to the primary inverter sections. Inthe exemplary embodiment, the separate current sources for theanti-parallel inverters are each configured to provide current levelsone fourth that of the primary inverter current sources. RPs 137 and138, and RNs 139 and 140, set the current levels at about one fourththat of RP 129 and RN 131, respectively, while RPs 141 and 142, and RNs143 and 144, are cascode configured to control the drain voltage of thecurrent setting devices. If the primary inverters are oscillating intandem, they must share current from current sources 129-130 and132-132, but even then have twice the current available as theanti-parallel inverter current sources. When the primary inverters areoscillating properly in opposing phase, the two inverter sections of thestage do not concurrently use the same current source.

FIG. 12B is an alternative implementation of a differential inverterstage that includes an anti-parallel inverter lock circuit. It differsfrom FIG. 12A primarily in the manner of current limiting the inverterstages. In FIG. 12A a single current source through FETs 129-130supplies the source current for both primary inverters to the sources ofFETs 121 and 125, while another single current source through FETs131-132 supplies the sink current for both primary inverters to thesources of FETs 122 and 126. Separate current sources provide source andsink current for the two phase opposition locking inverters of FETs133-134 and 135-136. In FIG. 12B, by contrast, one single current sourceconsisting of FETs 185-186 provides source current, and another singlecurrent source consisting of FETs 195-196 provides sink current, forboth phase opposition locking inverters comprising FETs 133-134 and135-136. Conversely, the separate current sources of FETs 181-182 and183-184 provide source current for the two primary inverters to FETs 121and 125, respectively; similarly, the separate current sources of FETs191-192 and 193-194 provide sink current for the two primary invertersto FETs 122 and 126, respectively.

The sizes of the inverters and associated current source transistorsalso differs between FIG. 12B and FIG. 12A. The width and length of thechannel is indicated in the figures by the numbers separated by aforward slash, indicating width/length, in microns. RP and RN FETs havethreshold voltages Vth of about 0.65 V and 0.7 V, respectively, whileVth for THP and THN FETs are about 0.95 V and 1.0 V. These are merelyguides; the size of the devices depends heavily on the particularfabrication process, as well as on the loading and other performancefactors of the oscillator. Also, although only two current limit schemesare set forth in FIGS. 12A and 12B, embodiments of such a differentialoscillator stage may employ many different current sourcing andanti-parallel phase opposition lock configurations without deviatingfrom the essential ideas set forth herein, and without exceeding thescope of claims set forth herein. FIG. 13 illustrates an exemplaryalternative means to provide phase opposition locking.

FIG. 13 illustrates a differential inverter stage that does not includeanti-parallel locking inverters, but instead illustrates an alternativewhereby capacitances may be cross coupled to ensure that thedifferential stages are 180 degrees out of phase. The basic inverterstage consists of the same numbered FETs, inputs and outputs 121-132 asillustrated in FIG. 12A. Capacitor 145 couples output outP 128 to inputin P 123, while capacitor 146 couples output outN 124 to input in N 127.The inverter pairs in a differential ring oscillator will have littletendency to synchronize in matching phase, and have some tendency tosynchronize in opposite phase because they then have access to the fullcurrent of a current source (FETs 129-130 or 131-132). It is for thisreason that the anti-parallel inverters of FIG. 12A require littlecurrent capacity. For the same reason, modest capacitance of about 200fF is adequate to ensure phase-opposite synchronization of the invertersof the inverter stage of FIG. 13. FIGS. 12 and 13 illustrate two out ofmany possible alternatives for ensuring phase opposition between theoutputs of a differential inverter stage. However, in some semiconductorfabrication processes, active anti-parallel inverter circuitry such asillustrated in FIG. 12A may require less semiconductor area than thecapacitors shown for the same purpose in FIG. 13.

Because they may function substantially identically to the primaryinverters of FIGS. 12A and 12B, the inverters of FIG. 13 are indicatedby identical reference designators. They consist of FET pairs 121, 122(producing outN 124 from in P 123) and 125, 126 (producing outP 128 fromin N 127). In FIG. 13 these two inverters shares a single source currentsource (FETs 129, 130) and a single sink current source (FETs 131, 132),identically as in FIG. 12A. In some circumstances it may be moresuitable to provide separate current source circuits for each of theinverters of FIG. 13 in the manner illustrated in FIG. 12B. Asillustrated there, source and sink current may be provided to the in Pto outN inverter via FET pairs 181, 182 and 191, 192, respectively, andto the in N to outP inverter via different FET pairs 183, 184 and 193,194, respectively.

Ring oscillators having an even number of inverter stages expand thedesign flexibility for ring oscillators, providing an additionalparameter to help control the operating frequency range. This isparticularly useful for embodiments required to include less than fiveinverter stages in order to produce a better output, because it triplesthe number of alternatives satisfying that requirement. An even numberof inverter stages may be employed by cross coupling the positive andnegative outputs of one stage to the negative and positive inputs,respectively, of the next stage. Because such cross coupling ensuresthat the outputs of each stage will be correctly out of phase, noinverter stage needs phase separating circuitry such as is illustratedin FIGS. 12 and 13. However, a ring oscillator having an even number ofstages does not unconditionally oscillate, so a provision for properstartup may be required.

FIG. 14 schematically illustrates a four stage differential ringoscillator 150 coupled to a startup circuit 160. The four differentialinverter stages 151-154 may each be configured as illustrated in FIG.13, except capacitors 145 and 146 are unnecessary. A capacitor 155 isdisposed between each differential inverter output and ground 156.Because the drive current of each inverter stage is constrained bycurrent limiting circuits for both source (RPs 129 and 130 in FIG. 13)and sink (RNs 131 and 132 in FIG. 13) currents, these capacitors 155 areable to smooth and shape the output to produce a substantially sine-likewaveform that is smooth and free of spurs and unwanted harmonic content,as described elsewhere herein.

Each of the outputs outN (124 in FIG. 13) of differential inverterstages 151-153 is coupled to the input in P (123 in FIG. 13) of thesubsequent stage. However, outN 157 of differential inverter 154 iscross coupled to in N (127 in FIG. 13) of differential inverter stage151. Similarly, each of the outputs outP (128 in FIG. 13) ofdifferential stages 151-153 is coupled to in N of the subsequent stage,except for outP 158 of inverter stage 154, which is cross coupled to inP of inverter stage 151.

The remainder of FIG. 14 schematically represents an example of astartup circuit 160 to ensure oscillation by differential ringoscillator 150. The inputs 161 and 162 to the startup circuit 160 may becoupled to the two outputs of any of the inverter stages 151-154.Although the outputs of the startup circuit 160, at the drains of FETs163 and 164, are connected to outputs outP 158 and outN 157 ofdifferential inverter stage 154, they could be connected instead, forexample, to the outputs of the differential inverter stage 151, as arethe inputs 161 and 162 of the startup circuit 160.

The startup circuit 160 is intended to identify a stable common modecondition in which both outputs of an inverter stages are stable, wheneither both are low, or both are high. Upon sensing that condition,startup circuit 160 will force the two outputs of each inverter stageinto opposite polarities (differential mode). Resistors 165-170 may allbe nominally around 2 MΩ or more, because the FET inputs to Schmitttriggers 171 and 172 draw practically no current. The large impedance ofresistors 165-170, in conjunction with FETs 173-176, cooperate withcapacitors 177 and 178 to increase noise immunity at the inputs ofSchmitt trigger devices 171 and 172.

In a first common mode condition both inputs 161 and 162 are low,biasing “on” the P-FETs 173 and 174 to gradually draw Schmitt trigger171 at least within a threshold voltage of the extremely low inputvoltages. While N-FETs 175 and 176 are biased off, the sources are atthe very low input voltages, and thus any positive charge on C 178 willbe drawn off by their combined channel leakage current, which willgreatly exceed any positive leakage from the FET input of Schmitttrigger 172. In a second common mode condition both inputs 161 and 162are high, biasing “on” the N-FETs 175 and 176 to slowly raise the inputof Schmitt trigger 172 by conduction via Rs 167, 168 and 170. Conductionto raise the voltage if Schmitt trigger 171 is limited to leakagethrough off-biased P-FETs 173 and 174. Though the channel leakage istiny, it will exceed leakage to ground through the insulated-gate FETSchmitt trigger input and through C 177. In the exemplary process,nearly all low voltage capacitors to ground are fabricated ascapacitor-connected depletion-type N (DN) FETs having negligible leakagecurrent. Thus, in both common mode conditions, the two Schmitt triggerswill eventually arrive at a common output voltage. Such same-polarityinputs will produce a low output from exclusive-or gate 179, directlybiasing on the first starter circuit output P-FET 163, and causinginverter 180 to bias on the second starter circuit output N-FET 164. Theactivated FETs 163 and 164 easily drive the outputs 158 and 157 ofdifferential inverter stage 154 to opposite supply rails because of thevery limited current drive capacity of the inverter outputs.

All inverter stage output pairs will have a stable and opposite polarityas long as output FETs 163 and 164 remain fully driven, including theoutput pair of inverter stage 151 that is coupled to inputs 161 and 162of startup circuit 160. Opposite (differential) polarity on the inputs161 and 162 will eventually cause the two Schmitt triggers to establishopposite states, resulting in the release of the startup circuit outputdrive such that oscillation of the ring will commence. In a firstdifferential alternative input 161 is low and 162 is high, so PFET 173is off and the enabled P-FET 174 can provide sufficient current frominput 162 to charge C 177 via Rs 166 and 169, eventually developing ahigh on the input to Schmitt trigger 171. Concurrently, N-FET 176 willbe off so that the forward biased N-FET 175 will gradually draw thevoltage of C 178 below the low threshold of Schmitt trigger 172 byconduction through R 167 and R 170. In the converse second differentialalternative input 161 is high and input 162 is low. N-FET 175 is off soforward biased N-FET 176 can gradually establish a low on the input ofSchmitt trigger 172. P-FET 174 is also off, permitting forward biasedP-FET 173 to raise the input of Schmitt trigger 171 toward the highinput 161 voltage via Rs 165 and 169.

As is seen, both possible opposite-polarity input conditions drive theinput of Schmitt trigger 171 toward high while driving the input ofSchmitt trigger 172 toward low. Those Schmitt trigger conditions will bereinforced during every half cycle during proper oscillation. The inputvalues approach each other only near the midpoint of the clock waveform,during which brief times the drive voltages are negligible.

CONCLUSION

The foregoing description illustrates exemplary implementations, andnovel features, of circuits and method for generating bias and auxiliarysupply voltages both quietly and efficiently. Many such voltages aregenerated by pumping charge via transfer capacitors without generatingexcessive electrical noise. Many features combine to produce the desiredresult, and are each described separately. Some features of apparatusand methods, which constitute the best mode of implementing quiet,efficient bias generation circuits and methods, are themselves novel andwidely useful. Consequently, the description set forth above necessarilydescribes a variety of distinct innovations.

The skilled person will understand that various omissions,substitutions, and changes in the form and details of each of themethods and apparatus illustrated may be made without departing from thescope of such method or apparatus. Because it is impractical to list allembodiments explicitly, it should be understood that each practicalcombination of features set forth above (or conveyed by the figures)that is suitable for embodying one of the apparatus or methodsconstitutes a distinct alternative embodiment of such apparatus ormethod. Moreover, each practical combination of equivalents of suchapparatus or method alternatives also constitutes an alternativeembodiment of the subject apparatus or methods. Therefore, the scope ofthe presented methods and apparatus should be judged only by referenceto the claims that are appended, as they may be amended during pendencyof any application for patent. The scope is not limited by featuresillustrated in exemplary embodiments set forth herein for the purpose ofillustrating inventive concepts, except insofar as such limitation isrecited in an appended claim.

The circuits illustrated and described herein are only exemplary, andshould be interpreted as equally describing such alternatives as may bereasonably seen to be analogous by a person of skill in the art, whetherby present knowledge common to such skilled persons, or in the future inview of unforeseen but readily-applied alternatives then known to suchskilled persons.

All variations coming within the meaning and range of equivalency of thevarious claim elements are embraced within the scope of thecorresponding claim. Each claim set forth below is intended to encompassany system or method that differs only insubstantially from the literallanguage of such claim, but only if such system or method is not anembodiment of the prior art. To this end, each element described in eachclaim should be construed as broadly as possible, and should beunderstood to encompass any equivalent to such element insofar aspossible but without also encompassing the prior art.

What is claimed is:
 1. An integrated operational transconductanceamplifier (“OTA”) having a variable-ratio current mirror differentialamplifier section comprising: a) a differential pair of FETs including afirst transistor M1 having a source S1 coupled to a source S2 of asecond transistor M2 and to a current source Ics, the two forming adifferential input pair of FETs for the OTA, the corresponding drains ofthe input pair forming a pair of differential current branches; b) anoutput voltage connection Vout+ coupled to one of the differentialcurrent branches; c) a variable ratio current mirror circuit, includingi) a current sensing circuit conducting a first current through one ofthe differential current branches; and ii) a current mirror reflectioncircuit that generates a second current substantially reflective of thefirst current, in the other differential current branch, the currentmirror reflection circuit comprising a third transistor arranged tooperate in parallel with a fourth transistor; and d) a current mirrorratio control circuit comprising a first control FET coupled in serieswith the fourth transistor of the current mirror reflection circuit, thefirst control FET including a gate configured as a mirror ratio controlnode to which is applied a voltage that is continuously variable forvarying a ratio between the first current conducted through the currentsensing circuit and the second current conducted through the currentmirror reflection circuit.
 2. The OTA of claim 1, wherein the ratio isvariable between 1:K and K:1 (K>1), and wherein the current mirrorreflection circuit is a single FET and the current sensing circuitcomprises at least one FET having drain and gate coupled together. 3.The OTA of claim 1, wherein the current sensing circuit is a single FETand the current mirror reflection circuit comprises a plurality of FETs,and wherein the first control FET is coupled to sources of the pluralityof current mirror reflection circuit FETs.
 4. The OTA of claim 1,wherein the variable-ratio current mirror differential amplifier sectionis a first such section, further comprising a second variable-ratiocurrent mirror differential amplifier section according to elements(a)-(d), the first and second amplifier sections having common inputscoupled to opposite differential inputs to produce inverting andnoninverting outputs, wherein the two variable-ratio current mirrors arecontrolled by the same mirror ratio control node.
 5. The OTA of claim 4,further comprising a circuit that couples the continuously variablevoltage that is generated at an output of the OTA to the mirror ratiocontrol node.
 6. The OTA of claim 4, further comprising a separateamplification circuit that generates the voltage that is continuouslyvariable based on a difference between the inverting and noninvertingoutputs.
 7. The OTA of claim 1, wherein the current mirror reflectioncircuit comprises three FETs coupled substantially in parallel, one ofthe parallel FETs having a source coupled to the first control FET. 8.An integrated operational transconductance amplifier (“OTA”) having avariable-ratio current mirror differential amplifier section comprising:a) a differential pair of FETs including a transistor M1 having a sourceS1 coupled to a source S2 of a second transistor M2 and to anapproximate current source Ics, the two forming a differential inputpair of FETs for the OTA, the corresponding drains of the input pairforming a pair of differential current branches; b) an output voltageconnection Vout+ coupled to one of the differential current branches;and c) a variable ratio current mirror circuit, including i) a currentsensing circuit that is a single FET, the current sensing circuitconducting current from one of the differential current branches thatgenerates a current control voltage reflective of the quantity ofcurrent thus conducted; ii) a current mirror reflection circuit thatgenerates a current substantially reflective of the current controlvoltage in the other differential current branch, the current mirrorreflection circuit comprising a plurality of FETs having drain and gatecoupled together; and iii) a circuit, proportionally controllable by asignal applied to a mirror ratio control node that controls a controlFET coupled between sources of the plurality of FETs of the mirrorreflection circuit within the OTA, which aids either (A) conductingcurrent of the current sensing circuit, or (B) producing currentreflective of the current control voltage, such that d) a ratio betweencurrent conducted by the current sensing circuit and the current mirrorreflection circuit is continuously variable over a range under controlof the signal applied to the mirror ratio control node.
 9. An integratedoperational transconductance amplifier (“OTA”) comprising: a firstvariable-ratio current mirror differential amplifier section,comprising: a) a differential pair of FETs including a transistor M1having a source S1 coupled to a source S2 of a second transistor M2 andto an approximate current source Ics, the two forming a differentialinput pair of FETs for the OTA, the corresponding drains of the inputpair forming a pair of differential current branches, b) an outputvoltage connection Vout+ coupled to one of the differential currentbranches; c) a variable ratio current mirror circuit, including i) acurrent sensing circuit conducting current from one of the differentialcurrent branches that generates a current control voltage reflective ofthe quantity of current thus conducted; ii) a current mirror reflectioncircuit that generates a current substantially reflective of the currentcontrol voltage in the other differential current branch; and iii) acircuit, proportionally controllable by a signal applied to a mirrorratio control node within the OTA, which aids either (A) conductingcurrent of the current sensing circuit, or (B) producing currentreflective of the current control voltage, such that d) a ratio betweencurrent conducted by the current sensing circuit and the current mirrorreflection circuit is continuously variable over a range under controlof the signal applied to the mirror ratio control node; and a secondvariable-ratio current mirror differential amplifier section accordingto elements (a)-(d), the first and second amplifier sections havingcommon inputs coupled to opposite differential input devices to producetwo outputs inverse from each other, wherein the two variable-ratiocurrent mirror circuits are controlled by the same mirror ratio controlnode.
 10. The OTA of claim 9, further comprising a circuit that couplesa signal substantially based on an output of the OTA to the mirror ratiocontrol node.
 11. The OTA of claim 9, further comprising a separateamplification circuit that couples a signal based on a differencebetween the inverting and noninverting outputs to the mirror ratiocontrol node.
 12. An integrated operational transconductance amplifier(“OTA”) having a variable-ratio current mirror differential amplifiersection comprising: a) a differential pair of FETs including atransistor M1 having a source S1 coupled to a source S2 of a secondtransistor M2 and to an approximate current source Ics, the two forminga differential input pair of FETs for the OTA, the corresponding drainsof the input pair forming a pair of differential current branches; b) anoutput voltage connection Vout+ coupled to one of the differentialcurrent branches; and c) a variable ratio current mirror circuit,including i) a current sensing circuit conducting current from one ofthe differential current branches that generates a current controlvoltage reflective of the quantity of current thus conducted; ii) acurrent mirror reflection circuit that generates a current substantiallyreflective of the current control voltage in the other differentialcurrent branch; and iii) a circuit comprising three FETs coupledsubstantially in parallel, one of the parallel FETs having a sourcecoupled to a source of another via a first control FET, and one of theparallel FETs having a drain coupled via a second control FET to drainsof the other two parallel FETs, the circuit proportionally controllableby a signal applied to a mirror ratio control node within the OTA, whichaids either (A) conducting current of the current sensing circuit, or(B) producing current reflective of the current control voltage, suchthat d) a ratio between current conducted by the current sensing circuitand the current mirror reflection circuit is continuously variable overa range under control of the signal applied to the mirror ratio controlnode.